Phase sector based RF signal decimation

ABSTRACT

Values representative of modulation signal components are extracted from a modulated signal. The modulated signal contains a modulation signal. A local clock signal is developed which correlates in time to the modulated signal and has a plurality of non-overlapping phase sectors per cycle. Signal values are acquired from the modulated signal, separately for at least one phase sector of one cycle of the local clock signal. At least two signal values from the same phase sector, but different clock cycles are combined to obtain at least one combined signal value representative of a modulation signal component.

RELATED APPLICATIONS

This application is a Continuation of co-pending U.S. application Ser.No. 13/651,259, filed Oct. 13, 2012.

BACKGROUND

There are many different modulation schemes used to modulate a RadioFrequency (RF) carrier or an Intermediate Frequency (IF) carrier with alower frequency modulation signal. There are different advantages anddisadvantages to each of the commonly used methods. There are also manydifferent wireless formats or standards used within the wireless productmarketplace today. These standards differ not only in modulationtechnique used, but also in the bandwidth utilized by the wirelesssystem, the RF bandwidth of a single channel, and the use, or not, ofvarious spread spectrum techniques, such as CDMA (Code Division MultipleAccess), frequency hoping techniques, or, most recently gaining inpopularity, OFDM (Orthogonal Frequency Division Multiplexing). Anidealized goal of a Software Defined Radio (SDR) system is to be able totransmit and receive using any of these techniques and to be able toswitch between them by merely changing the software code running on suchan SDR system.

Cognitive Radio Systems, as currently envisioned within the R&Dcommunity, include plans for flexible transceiver's, which can adjust toband utilization variations as needed, changing frequency bands,modulation techniques, and transmission bandwidths as required to makebest use of the current RF environment. This discussion continues, yetcurrently there is not even a cost effective or efficient method toimplement a fully flexible SDR.

Given the lack of fully flexible SDR technology, each of the manydiffering modulation techniques and wireless standards have historicallyrequired different customized analog front-end receiver blocks andcustomized back-end analog transmitter blocks for each bandwidth,modulation technique, or wireless standard accommodated. The concept ofSDR has often been put forth with the promise of a single circuit blockthat could, under software control, be able to operate and providecompetitive performance, while working with any of the current wirelesstransmission schemes. Yet this promise remains unfulfilled.

The roadblocks to achieving fully flexible SDR solutions have been many.The use of IF stages, creating difficult to manage spurs, and imagefrequency artifacts, also create overly complex matrices of usabilitylimitations. Each combination of center frequency, bandwidth, andmodulation scheme has required specific design attention despite atheoretically programmable feature selection. Quadrature modulation andall schemes which make use of phase variation add a great deal ofcomplexity to both reception and transmission. Maintaining orthogonalityand minimizing mismatches between the I and Q channels are ongoingchallenges. Conventional zero IF approaches to the SDR challenge attemptto simplify the IF complexities, but compound the I/Q mismatch andorthogonality issues and typically degrade noise performance as well.Real solutions to the SDR challenge have seemed perennially imminent,while remaining ethereal.

DESCRIPTION OF THE DRAWINGS

FIG. 1 is a system block diagram showing one embodiment of the presentinvention implemented in a phase sector based software defined radio.

FIG. 2 is a timing signals chart for quadrature and phase sectorprocessing.

FIG. 3 a is a diagram illustrating one embodiment of a phase sectordistributed integrator of the present invention.

FIG. 3 b is a timing diagram for the diagram of FIG. 3 a.

FIG. 4 is a diagram illustrating one embodiment of alternating blocks asphase sector distributed integrators with a charge responding A to D.

FIGS. 5 a-b are a flow chart showing one embodiment of the presentinvention method for extracting values representative of modulationsignal components from a modulated signal.

FIG. 6 is a flow chart showing one embodiment of a method foraccumulating a modulated signal with a capacitive device.

FIG. 7 is a flow chart showing one embodiment of accumulating andcombining accumulated values.

FIG. 8 is a flow chart showing an alternate embodiment of accumulatingand combining accumulated values.

FIG. 9 is a flow chart showing another embodiment of accumulating andcombining accumulated values.

FIG. 10 is a flow chart showing one embodiment of a method forcontinuously accumulating charge.

FIG. 11 is a flow chart showing a method for using the alternatingblocks of FIG. 4.

FIG. 12 is a flow chart showing one embodiment for the method of thepresent invention including a transmit mode.

FIG. 13 is a flow chart illustrating one embodiment for altering a localclock signal.

FIG. 14 is a diagram illustrating one embodiment of alternating blocksas phase sector correlated decimation filters, combining sampled valuesby accumulation, and passing accumulated values to a charge responding Ato D.

DETAILED DESCRIPTION

There are many common terms used in the RF industry which are generallyunderstood to have consistent meanings, but which can, by usage or byapplication, have slightly varying scope or specific meaning. Forclarity, there are three terms which will now be defined veryspecifically, which in some cases will be used in place of more commonusage meanings or terms, other such specific terms will be describedupon initial use. A modulated signal is one such term, carrier signal isanother, and modulation signal is another. A modulated signal is asignal which includes information representative of a modulation signal,which is superimposed, encoded, or modulated onto a carrier signal tobecome the modulated signal, and from which the original modulationsignal can be recovered by some means. Generally, most if not all RadioFrequency (RF) signals are modulated signals, as are all IntermediateFrequency (IF) signals. The carrier signal is a single frequencysinusoidal waveform, generally higher in frequency than any of thespectral content of the modulation signal. The modulation signal is asignal superimposed on a carrier using any of a wide variety ofmodulation techniques common in the art.

Developing a more ideal receiver solution for SDR has resulted invarious novel methods, blocks, circuits, and systems. Although thepresent invention is not limited in scope to SDR applications. FIG. 1,shows a top level block diagram of a Phase Sector Based Software DefinedRadio, (PSB-SDR) system 2, based on these developments. The RF2D block4, a novel development, is an input block that receives a wide-bandmodulated signal from a wide-band amplifier 8, or perhaps directly froman antenna, and outputs to a DSP block 10 a wide-band stream of datawhere each piece of data is representative of a modulation component andwhere the data is phase sector correlated. A modulation signal componenthere is any quantity, value, or signal, which when combined with othermodulation signal components, can form a representation of a modulationsignal. A representation of a modulation signal thus formed is thenreferred to as a reconstructed modulation signal.

To understand phase sector correlation, phase sector must first beunderstood and defined. A phase sector represents a span of the phase ofa local clock signal, which remains essentially constant from one clockcycle to the next in the phase angle of the local clock at which itbegins and in the phase angle of the local clock at which it ends. Eachcycle of the local clock signal has multiple phase sectors. While it isnot necessary that phase sectors be contiguous, where every possiblephase angle of the local clock is thereby included in one or anotherphase sector; it is desirable that phase sectors remain non-overlappingso that no two phase sectors both include any one phase angle of thelocal clock. Generally, it is intended that phase sectors be nearlycontiguous, so that every phase be included in one or another phasesector, except for transition phases at the beginning or end of a phasesector. However, it is possible to have phase sectors which havesignificant gaps between them, yet they should never overlap.

The local clock signal is an approximately constant frequency signal,generated by a local oscillator block 12, or timing system, where localmeans within the context of the receiver system, which during the normaloperation of receiving a signal is intended to be at the same frequencyand in a constant phase relative to the carrier signal used to constructthe modulated signal input to the RF2D block 4. Generally, the actualcarrier is generated at the transmitter, and the pure carrier signalitself is therefore not generally available at the receiver.Furthermore, the carrier signal is not necessarily even present withinthe modulated signal, depending on modulation and transmission scheme.There are various methods used to establish the correct frequency andphase for a local clock. Most methods use some form or other of either aphase or a frequency locked loop system. Such phase lock is generallyachieved by first defining some window of time, established by timingformat, during which the modulated signal can be relied upon to berepresentative of the carrier frequency, and to be at some knownreference phase. However, precisely how this works can varysubstantially from one system to another. The system herein described isan SDR system intended to be capable of receiving essentially anytransmitted signal. It is therefore necessary for the system to becapable of all methods of frequency or phase lock.

In the most general case, the most that can be relied upon is that thelocal clock, when in proper frequency and phase relationship relative tothe source carrier, can generally be said to be correlated in time tothe modulated signal. It is important to note however, that this is notalways a strong mathematical correlation in the strictest sense. Wherethe carrier is suppressed (not transmitted) and if the modulation factoris high (a statistically high percentage of the modulation range isutilized) this means there may not be very much of the carrier frequencycontained in the modulated signal, such that the mathematicalcorrelation may not be very high at all. In such cases, as long as thesignal content of the modulated signal can still effectively bedemodulated by any means, using the local clock at a frequency near orin the band of the modulated signal, then the local clock signal canstill be considered as correlated in time with the modulated signal.

The term correlation is used far more strictly and quite differently forthe term phase sector correlation. Here, unlike with the local clockwhose correlation in time with the modulated signal might be quite lowmathematically, yet still be considered correlated, phase sectorcorrelation is intended to indicate a very high level of mathematicalcorrelation, virtually 100%. Each phase sector amounts to a window oftime during which the modulated signal gets acquired or captured, withthe value captured becoming associated only with the phase sector activeduring the time of capture. The fact that phase sectors arenon-overlapping ensures that any instantaneous time value of the inputsignal gets included in only one phase sector. This maintains theindependence of the modulated signal values so captured and thesignificance of the phase sectors as separate from one another. Signaltransmitted or received within a phase sectors span remains isolated andseparated from signal transmitted or received during other phasesectors.

Phase sector correlation can now be understood and signal or data can beunderstood to be phase sector correlated whenever all data or signalcaptured during any one phase sector is collected or captured and keptseparate from data or signal captured or collected during any otherphase sector. Wherever this separation is maintained, the data or signalcan be said to be phase sector correlated. Again, this correlation is atype of correlation where a high mathematical correlation is importantin order for it to be phase sector correlated. In fact, phase sectorcorrelation indicates a case where full correlation is virtually assuredby design or by definition.

The data stream output of the RF2D converter block 4, RF2Dout, has anadjustable data rate, f_RF2Dout, adjustable under the control of the DSP10 and generally chosen to be fast enough to at least provide Nyquistrate data relative to the widest band information present in theincoming signal. In this way, no loss of bandwidth in the data or signalreceived occurs within the RF2D block 10. It is important to notehowever, that it would be possible to allow a reduction in bandwidth, ifso desired, within the RF2D block merely by selecting an RF2Dout speedslower than the Nyquist rate relative to the widest band informationpresent in the incoming signal.

The first processing block within the RF2D block 4, is a discrete timeprocessing block called Phase Sector Correlated Capturing (PSCC) 14,where the modulated signal input to the RF2D block 4 is broken up intodiscrete time segments and thereby sampled or acquired according tophase sector and then processed without yet being converted to a digitalor binary representation of the signal. As such, the signal in thisfirst stage still has full analog signal resolution, where the effectiveresolution is limited only by a noise floor present. This noise floor isformed as a combination of local circuit processing noise and noisepresent in the incoming signal. In one embodiment, this discrete timeblock also includes multiple channels of low-pass filters, which actlike discrete time decimation filters on the input data. There is onefilter for each phase sector. Each of these phase sector correlateddecimation filters has a decimation rate that is variable depending onthe number of local clock cycles over which the sampled values areacquired. The simplest discrete time filter for this application isimplemented by simply adding up (or “accumulating”) the captured signalvalues (or “sampled values”), separately by phase sector, from one cycleto the next. In this way, each value captured during any one phasesector of one cycle of the clock is simply added together with signalvalues captured during the same phase sector of subsequent clock cycles.

This low pass filtering is most simply achieved by the adding up, or theaccumulation, of the phase sector correlated values from one clock cycleto the next. This most typically accumulates one additional value perphase sector, once for every local clock cycle that occurs during theentirety of one full RF2Dout cycle period. Once accumulated, thisresults in one value passed to the DSP for each phase sector per RF2Doutcycle period. Since this occurs within a discrete time analog resolutionblock, the noise which is typically random or white in nature would tendto add up stochastically, whereas the signal tends to add up linearly.This yields a profound advantage for this methodology. This means thesignal to noise ratio tends to improve as the signal is accumulated, bya factor roughly equal to the square root of the down-conversion factor,where the down-conversion factor is the RF or local clock rate dividedby the RF2Dout rate, f_RF2Dout. This accumulation of values from oneclock cycle to values accumulated from other clock cycles can generallybe done with any value combining system, and is probably most simplyimplemented using switched capacitor techniques.

Another option, rather than down-conversion, is to pass the signalcaptured for each phase sector to the A to D 16 at full speed, one valueper phase sector per local clock cycle. This amounts to f_RF2Dout=Fclk,where Fclk is the frequency of the local clock, which is most typicallyequal in frequency to the carrier frequency of the modulated signal.This would require that the DSP 10 process values coming in from the Ato D 16 at a rate equal to the number of phase sectors per clock cycletimes the Fclk rate. For a quadrature signal, that would be at leastfour times the Fclk rate. Depending upon the carrier frequency involved,that can become a large amount of individual pieces of data processedvery quickly, in effect using up a great deal of the DSP block'sprocessing power. This can also generate quite a speed challenge for theA to D block 16. Under some signal conditions this might be thepreferred processing method, particularly where Fclk is not too high,but in most conditions, it is more desirable to reduce the input signalrate to the DSP 10, using the down-conversion option. In fact, in manysystems this down-conversion capability is a fundamental enabler,without which, such a system simply cannot be made to work fast enough.

For a flexible and programmable system, as this PSB-SDR system 2 isintended to be, it is generally desirable to be able to capture datanearly continuously. In order to avoid regular dumping intervals duringwhich the input signal is ignored, the RF2D block 4, in one embodiment,includes two identical discrete time processing input blocks, onereceiving input values, while the other is dumping its values to asubsequent block, usually the A to D block 16. FIG. 4 illustrates oneembodiment of these two blocks 18, 20 (accumulation blocks) for a phasesector integration capturing system, described below. Each of these twoblocks also has their value reinitialized during its dumping period,reinitialized to a level which the A to D 16 would receive and interpretas a zero value. This occurs after it has dumped its value to the A to D16 but before it is reconnected to new input for its subsequentacquisition phase. In this embodiment, these two blocks are includedwithin in the Phase Sector Correlated Capturing block 14 of FIG. 1.These two blocks then alternate roles on alternate RF2Dout cycles. Thisallows the system to collect signal for every local clock cycle.Ultimately, this provides a better signal to noise ratio of theresulting recovered modulation signal, for any given modulated signalinput, than a scheme which ignores any number of cycles of modulatedsignal input. This also potentially avoids, or at least minimizespotential aliasing problems, and the complexities of managing adjustableanti-aliasing filters.

Regardless of the carrier frequency of the incoming modulated signal,regardless of its bandwidth, regardless of the modulation schemeutilized, and regardless of the information the modulation contains;there is some selection of the Fclk/RF2Dout down-conversion factor, thenumber of phase sectors per clock cycle, and the selection of localclock frequency; such that the resulting digitized phase sectorcorrelated values can be passed to the DSP block 10, containing theinformation necessary to fully decode, demodulate, filter, and presentdata in whatever form most desirable, for any RF signal which can betransmitted, and to do so in a way which provides the best signal tonoise ratio possible. Because of this, this system is capable of thefull flexibility envisioned in the original concept of SDR, as noconventional or prior art circuits or systems have been.

It is instructive to consider one specific type of modulation, to seehow PSCC provides benefits for a specific case. Quadrature modulation isused for many different systems, 8QAM, 16QAM, and 32QAM to name aspecific few, and countless others. Quadrature modulation is also key toconsider because the quadrature relationship of the I and Q signals addsthe significance of phase variation, as well as generally addingcomplexity to the demodulation process. A quadrature modulated signal ismost simply processed using PSCC, with four phase sectors per localclock cycle. FIG. 2 includes waveforms, which show the timing for eachof four equally sized phase sectors at (a), (b), (c), and (d). In orderto have these timing relationships, a phase lock condition between thecarrier that formed the modulated signal and the local clock must exist.It is simplest to merely assume that this condition does exist for now.Such a phase lock condition can be accomplished through conventionalmeans, using a phase locked loop, but a much preferable enhanced methodfor achieving this phase lock flexibly is also discussed further below.

Simple impulse sampling may not provide the best signal to noise ratio,but is the most common method of acquiring a discrete timerepresentation of a signal, and therefore an interesting case toconsider. Applying PSCC to simple impulse sampling results in atechnique described below as Phase Sector Impulse Sampling, or PSISImpulse sampling must be done at regular intervals. This, in combinationwith phase sector correlation, requires that each phase sector includean equal number of evenly time-spaced samples. For a four phase sectoror quadrature case, this necessitates an integer multiple of foursamples per Fclk cycle. The simplest case is again the best to consider,which is just the single sample per phase sector case. The onlyremaining choice then is where to phase the impulse sample within thephase sector. Aligning each impulse sample time with the center of itsphase sector, becomes the case to consider, for reasons which becomemore apparent as phase locking is considered, as described in greaterdetail below.

Phase Sector Impulse Sampling, PSIS operates by first acquiring asampled value using an impulse sampling method. This is done for thePSIS system shown in FIG. 14 for each of four phase sectors, during thetime when each of the sampling switches, SWa, SWb, SWc, SWd are closed.During each of these respective intervals, the sample capacitors Ca, Cb,Cc, Cd, each acquire whatever charge is necessary so that the voltage onthe sample capacitor closely matches the input signal voltage by the endof the sample interval. This acquired charge then becomes the sampledvalue. It is instructive to note, that generally, with the use ofswitch-cap circuits, it is important to consider, and to factor out ofthe signal, any variation in offset voltages present at the input of opamps. These techniques are well understood in the art, so forsimplicity, these details have been left out of FIG. 14. Any of theseoffset cancellation techniques could be used in conjunction with thiscircuit without altering its functionality pertaining to the conceptsherein presented.

Once a sampled value is thus acquired on each of capacitors, thesampling switch is then shut off. Each of the respective complementswitches, SWa_BAR, SWb_BAR, SWc_BAR, SWd_BAR then transfers the chargeon the sample capacitor, the sampled value, to the accumulationcapacitor of either the block 1 or the block 2 decimation filter,whichever one is in capture mode. Note that in the drawing SWa_BAR isdesignated by the letters SWa with a line above it to designate the BARnotation, indicating that the timing for SWa_BAR is the inverse of thetiming for SWa. It is also important to note, that the timing for eachof the complement switches is non-overlapping with that of the samplingswitches, as is typical for most switched capacitor circuits, so that nosampling switch and its respective complement switch are ever on at thesame time. The timing of the sampling switches relative to each othercan be seen in FIG. 3 b. However, it is important to note, that thetiming of signals a b c and d of FIG. 2 are arranged to show theintervals of each phase sector, but these are NOT phased the same as theswitch control signals for the PSIS system. The effective moment ofimpulse sampling occurs at the end of the sampling interval. This shouldoccur at the moments of impulse sampling shown in FIG. 2 e, which aspreviously described, should occur at the center of a phase sector, asshown in FIG. 2, for the case where only one sample is captured perphase sector.

Each accumulation type decimation filter of FIG. 14 then continues toaccumulate sample values by phase sector during one entire cycle of thef_RF2Dout clock, At the end of that clock period the decimation filterwhich had been capturing, switches to transfer mode, and begins passingthe accumulated values for each phase sector to the A to D, one value ata time. While the other decimation filter block is then put into capturemode, and begins accumulating sampled values.

There are various uses of capacitors made throughout the circuits,systems, and blocks of the PSB-SDR 2. Each capacitor as described hereand throughout this writing could be replaced with a capacitive device.A capacitive device, being any device or plurality of devices generallyhaving two or more terminals, which has capacitance between twoterminals such that it behaves like a conventional capacitor, in amanner sufficient so as to allow the circuit or system to behaveapproximately as it would if a capacitor were in its place, generallyhaving an amount of charge stored on the device equal to the voltageacross the device multiplied by its capacitance. For simplicity, theterm capacitor will be generally used, with the understanding that itcould be replaced by any capacitive device.

FIG. 2 (e) shows an incoming modulated signal, having an all in-phase(I) signal, which means that the I component is maximized, and thequadrature (Q) component is zero. While a real RF signal would havephase variation, there are only three full cycles of the modulatedsignal shown. The bandwidth limitations on most RF signals, would notallow the modulated signal to vary significantly in phase over threecycles, so the pure sine wave shown for FIG. 2 (e) is reasonablyrealistic over the time window shown. For this case, the impulse samplesare also shown in FIG. 2 (e), where there are four impulse samples percycle of the modulated signal. From left to right, these impulse samplesshown in FIG. 2 (e) are for phase sectors A, B, C, D, A, B, C, D, A, B,C, D. PSCC allows for multiple impulse samples to be accumulated overmultiple Fclk cycles, as long as the accumulation is done separately foreach phase sector, yielding a separate accumulated value for each phasesector, and so that any values accumulated are captured within the samephase sector, although probably over multiple cycles. This accumulation,for the PSIS case, can be accomplished with the addition of aswitched-capacitor circuit designed for this purpose, FIG. 14 shows twosuch blocks. Here each switched capacitor filter provides an analogdecimation filter, separately for each phase sector, accumulating thesum of the values captured on a capacitor, for later transfer to the Ato D block 16. For the signal state shown in FIG. 2 (e), all of thephase sector A values would be accumulated into a single value alongwith those from subsequent cycles, for however many local clock cyclesare included in a single RF2Dout cycle. Accumulation is the simplestapproach, but a more complex switched capacitor filter could also beused here to achieve steeper band response features.

From FIG. 2 (e), it is plain to see that all of the phase sector Avalues are identical, as are all of the phase sectors B, C, and D valuesrespectively. This is a direct result of the bandwidth being too narrowfor the phase of the modulated signal to vary too quickly, incombination with the modulated signal having the same frequency as theFclk. As described above, the down-conversion ratio of Fclk/RF2Doutwould normally be chosen so that the bandwidth of the resulting RF2Doutput data would not be significantly reduced, which means generally,that there would not be much variation in the values of all of the Aphase sector values over the time of their accumulation. Naturally, thisalso applies to the B, C, and D values respectively.

Each of the values for phase sectors A, B, C, and D, whether combinedover multiple Fclk cycles or not, gets transferred and converted to adigital value by the same A to D converter 16, with resulting digitalvalues then made available to the DSP 10, while their phase sectorcorrelation also is provided to the DSP 10, so that each value is stillidentified by the A to D as data resulting from phase sector A, B, C orD. Mixing of these phase sector correlated values can then be easilyperformed by the DSP. The I or in-phase signal can be formed by addingdata from the A and B phase sectors and subtracting the data from the Cand D phase sectors, or I=A+B−C−D, where A, B, C, and D now representdigital values collected during each of their respective phase sectors,possibly over multiple Fclk cycles. The Q or quadrature signal issimilarly developed by combining the phase sector values as follows,Q=B+C−A−D. In this way, each of these combinations yields one value for1 and one value for Q per cycle of RF2Dout.

The conventional approach to performing a phase locked loop (PLL)function is to use a phase comparator that compares the phase of areceived modulated signal to that of the local clock, usually during apredefined reference period discerned from receiving a signal with theproper timing format. The phase comparator most conventionally uses afour quadrant multiplier or an exclusive-OR block that essentiallyimplements a binary type multiplication on the two input signals, whereone is the local clock and the other is the received modulated signal,where the multiplier is only on during a predefined phase referenceperiod. When not on, the output of this phase comparator is disabled anda low pass filter, most often just a single capacitor, holds its voltageunaltered until the next predefined reference period occurs. This typeof phase comparator tends to cause a phase lock loop to lock with thetwo inputs to the phase comparator in quadrature with one another thatis 90 degrees out of phase, which results in an equal period of chargingthe hold capacitor as discharging it per local clock cycle, during thepredefined reference period. The output of this phase comparator isoften that of a charge-pump, charging or discharging the hold capacitorin response to the relative phase of the local clock and referenceperiod signal. The voltage on the capacitor then typically becomes thecontrol voltage input for a voltage controlled oscillator.

This same functionality can be achieved, using phase sector correlatedcapturing as applied to a quadrature modulation case, by employing adigital low pass filter technique on the post conversion mixed Q signal,developed within the DSP, or, alternately, by acting on a separatelydeveloped control signal having control signal values developed withinthe DSP from accumulated values previously discussed. This low passfiltering would then be done to obtain a control value or voltage to beapplied as a control input to the local oscillator, thereby controllingits frequency, as with a convention Voltage Controlled Oscillator,(VCO), and thereby forming a PLL 10. Alternately, some of the low passfunction could be reserved for an analog final stage, which could be acharge pump type design, either charging or discharging a holdcapacitor, similar to that of a conventional PLL phase comparator outputand VCO control input, but with the signal driving it being a digitallylow pass filtered version of the Q signal. This digital low pass and/orcharge pump could then be programmed, under the control of the DSP 10,to be enabled only during the correct reference period, thereby againcreating an effective sample hold on the phase comparator output. Thevoltage on the hold capacitor is then applied to a VCO, and so controlsthe oscillator frequency used to develop the local clock, and the timingof the four phase sectors. This now essentially forms a fullyprogrammable PLL. The programming for this could be changed, the precisefilter response of the low-pass could be changed as with the rest of thefeatures of this SDR system 2, to accommodate any signal/modulationformat, now including the timing details of the desired phase referenceperiod. The programming could also be changed in response to signalconditions, by having a control signal evaluator, probably formed merelyby program steps operating within the DSP, which can evaluate thecontrol signal described above, before it is low pass filtered, or anyof the other signals otherwise developed within the DSP, comparing thesignals to any number of characterization metrics, many determined bymere mathematical processing of the control signal, to determine whatthe low pass filter characteristics should be, or to determine how thecontrol signal is processed in developing the control value or voltageused to control the VCO.

Mixing the accumulated values after any down-conversion desired, andafter A to D conversion has taken place, constitutes a novel methodologycalled Post Conversion Mixing. This Post Conversion Mixing isadvantageous in every way, as it provides for virtually no mismatchbetween either the I and Q channels, nor between either of those and thephase detection mixing traditionally used to provide the control voltageto control the voltage controlled local oscillator of the PLL. All ofthis mixing is now done post down-conversion and post analog to digitalconversion, and is therefore acting on data which has been processedthrough a single analog block and a single analog to digital conversionprocess. I and Q are now formed by mathematically mixing identical data.This leaves essentially no place for I/O mismatch to occur within thereceiver. This is a major advancement over conventional techniques.

Previously, a method implementing PSCC using standard impulse samplingtechniques was described, PSIS. An alternative is to use Phase SectorIntegration Capturing (PSIC), a novel technique capable of improvednoise performance and capable of continuous signal capturing, a furtherenhancement to be described in detail below.

The most straight-forward or simplest way of implementing Phase SectorIntegration Capturing, is to consider FIG. 3 a, a phase sectordistributed integrator 22. For FIG. 3 a, this block 22 and its signaltiming is arranged to show a specific case of Phase Sector IntegrationCapturing, and a special case of Phase Sector Distributed Integrator,where the system is set up to perform quadrature detection. Similarly tothe description above considering the use of impulse sampling, thisarrangement includes a specific selection where there are four phasesectors per local clock cycle, and where the phase relationship betweenthe local clock and the incoming modulated signal is phase locked, justas in the impulse sampling case, in a manner and with a static phaserelationships consistent with that of traditional quadrature signaldetection methods.

The input signal, a modulated signal, which might be received directlyfrom an antenna, or from a very wide-band amplified version of signalpresent at the antenna, or perhaps an attenuated output of atransmitter, is applied to one side of a resistor Rin, where the otherside of the resistor is connected to the inverting input of an idealop-amp 24. The positive input 26 to the op-amp is connected to a DCvoltage reference level, which might be ground, and is certainly aneffective AC ground. The output 28 of the op-amp is connected to fourswitches, SWa, SWb, SWc, SWd (for the quadrature case), where only oneof the switches is closed or on at any one time. The other side of eachswitch is connected to one side of a capacitor, Ca, Cb, Cc, Cd, with theother side of the capacitor Ca, Cb, Cc, Cd connected back to theinverting input 30. Each switch SWa, SWb, SWc, SWd is connected to itsown capacitor Ca, Cb, Cc, Cd. In this way, because of the switches SWa,SWb, SWc, SWd, only one of the capacitors Ca, Cb, Cc, Cd is connected atany one time. While one of the switches SWa, SWb, SWc, SWd is closed,the op-amp 24 acts on the capacitor Ca, Cb, Cc, Cd through the switchSWa, SWb, SWc, SWd, providing whatever voltage is required to keep theinverting input of the op-amp 24 essentially and approximately equal tothe positive input 26 of the op-amp 24. Since the inverting input nodeis then maintained at a constant voltage, in this case at ground, anysignal present at the input side of the resistor Rin, is converted intoa current through the resistor Rin. This current has no where to goexcept through the capacitor Ca, Cd, Cc, Cd having its related switchSWa, SWb, SWc, SWd on. In this way each capacitor Ca, Cd, Cc, Cd, ischarged with a current that is a replica of the input signal, andthereby the charge on the capacitor Ca, Cd, Cc, Cd becomes proportionalto, and a replica of, the integral of the input signal during the periodwhen its associated switch SWa, SWb, SWc, SWd is on. In this way, thisblock 22 becomes essentially a phase sector accumulator, accumulating avalue, in this case charge, over each non-overlapping, timing systemdeveloped phase sector.

Each of the four switches SWa, SWb, SWc, SWd is then sequentially turnedon, after the other switches SWa, SWb, SWc, SWd are all off, so thatthere is no overlapping time period where more than one switch is on.This prevents charge representing signal, which has correctly beenacquired onto one capacitor Ca, Cd, Cc, Cd, from being altered by signalduring a different phase of clock, or by charge on an alternatecapacitor Ca, Cd, Cc, Cd. The timing of the turning on of each of theswitches SWa, SWb, SWc, SWd is sequentially arranged, so that any oneswitches SWa, SWb, SWc, SWd period of on time, always follows the onperiod of the same other switch SWa, SWb, SWc, SWd. So, each switch andcapacitor combination A, B, C, and D, is named so that the time whenswitch SWa is on always follows when switch SWd is on, SWb alwaysfollows SWa, SWc always follows SWb, and SWd always follows SWc, thesefour periods together complete one full cycle of the local clock.

As described above, the circuit from FIG. 3 a operates so as tointegrate the input signal current, and thereby store a charge on eachof the capacitors, which is proportional to the integral of the inputsignal over the period where the switch associated with that capacitoris on. This then results in charges, referred to as Qa, Qb, Qc, and Qd,which are the charges representative of accumulated values, specificallyanalog accumulated values, on each of the four capacitors Ca, Cb, Cc, Cdof FIG. 3 a, during periods A, B, C, and D, respectively. FIG. 3 b showsthe timing of SWa, SWb, SWc, and SWd.

After as few as one full cycle (or perhaps after multiple cycles) eachof the capacitors Ca, Cb, Cc, Cd, which have now accumulated a signalrelated charge as an accumulated value, are then dumped into a chargeresponding A to D converter 16, where a digitized accumulated value isdeveloped, which is directly proportional to the amount of chargeaccumulated on each capacitor Ca, Cb, Cc, Cd. A separate A to D outputvalue is acquired for each phase sector value, one for the chargeaccumulated during each of quadrature phase sectors A, B, C, and D.

If the conversion to digital is done at the end of every local clockcycle, there is one digital value for each quadrature phase A, B, C, andD, for each local clock cycle. However, if the integration continuesover multiple local clock cycles, then there is only one digital valuegenerated for each quadrature phase A, B, C, and D, for eachmultiplicity of local clock cycles. The number of cycles over which theintegration continues before executing an A to D on the integratedvalues, becomes the down conversion factor. If the integration continuesover five cycles, then the accumulated value converted by the A to D 16is first integrated for each quadrature phase, over five cycles, and thevalue converted to digital is effectively an average of the inputsignals integration by phase sector, over the five cycles. Thiseffectively executes a down conversion by a factor of five. This meansthe A to D 16 does not have to execute an A to D nearly as quickly.Furthermore, as before with the impulse sampled case, the data isaveraged, so any random noise is reduced, while the signal in effect,gets larger because of the ongoing integration. None of the desired passband data is lost unless the down-conversion factor becomes large enoughto result in a restriction of the bandwidth to a value that is narrowerthan that of the data contained in the modulation of the modulatedsignal. Ideally, the down-conversion factor is chosen so as to make thebandwidth just large enough to avoid significantly limiting thebandwidth of the data. The exact bandwidth chosen would likely be chosento provide the best signal to noise ratio of the resulting data.

If a local clock circuit which generates the full cycle described above,is synchronous (at the same frequency) with the carrier frequency of amodulated signal present at the input of the circuit of FIG. 3 a and ifthe static phase relationship between the local clock and incomingmodulated signal is correctly set, the analog accumulated valuerepresented by the charge accumulated on each of the capacitors Ca, Cb,Cc, Cd, after one full cycle, could then be selectively combinedtogether in the correct polarities, and thereby reconstruct one RFcycle's worth of demodulated I and Q base-band signals, for that cycle.

Phase Sector Integration Capturing can be used in this way, to achievemodulation signal values, using analog discrete time methods.Selectively combining each of the capacitors charge values, with thecorrect polarity selection for each, amounts to the mixing function, andthereby reconstructs analog discrete time in-phase or quadrature signalvalues. This combining can be accomplished using any accumulated valuecombining system. Where the processing block is analog, the values areusually charges, and the combining is most commonly implemented usingswitched-capacitor techniques. Where the combining is accomplished usingdigital values, the accumulated values are just digital values in activememory, and the accumulated value combining system generally becomes amathematical step executed by a program running on within a DSP block.Considering the case where analog modulation signal values are formedbefore digitization, this is done for each clock cycle, according toQi=Qa+Qb−Qc−Qd, and Qq=Qb+Qc−Qa−Qd, where the charges Qi and Qq soformed are now representative of modulation signal values, analogmodulation signal values, with Qi as a charge representing an in-phasemodulation value, a single cycles reconstruction of the in-phasemodulation, and with Qq representing a quadrature modulation value, asingle cycles reconstruction of the quadrature modulation.

As described earlier however, there is great advantage in performingthis mixing function in a post conversion manner instead. The point hereis that these analog accumulated values contain the significance of thequadrature and in-phase signal components, and that this is onealternate and novel way that this Phase Sector Integration Capturingtechnique could be used. If this system were used this way, wherediscrete time analog modulation values are formed, it is also possibleto then further combine these already mixed analog values, using amodulation value combining system, which would combine modulation valuesfrom multiple clock cycles, inherently creating a discrete time filter.This modulation value combining system could also be implemented withinthe DSP block, merely by performing the correct mathematical operationson digitized modulation values. Where implemented, this modulation valuecombining system would most typically be used to create a low pass,discrete time, or digital decimation filter.

The distributive property of multiplication over addition also appliesover accumulation and suggests that if charge is accumulated first byphase sector and the accumulated values are then combined together inthe correct polarity, that the resultant can be no different than if thevalues are first multiplied by the correct polarity and then accumulatedall at once. Conventional mixing is achieved by multiplying themodulated signal by the correct polarity, and then accumulating all atonce. This is necessarily done separately for I and Q, once multiplyingby the in-phase clock, FIG. 2 (f), and once by the quadrature clock,FIG. 2 (g). Each of these mixers outputs then get separately low passfiltered, similar to integration over its reject band, which isessentially accumulated all at once, without regard to phase sectors.Phase Sector Integrated Capturing can achieve the same resultants, butin a way that provides much greater flexibility by accumulating by phasesector first, then combining in the correct polarity to achievemodulation signal values.

For full flexibility, selective combining of accumulated values can beperformed on digital accumulated values, after digitizing the analogaccumulated values. The option of reconstructing the modulation signalvalue before digitization, by selectively combining the analogaccumulated values, has already been described. Alternately, under somesignal conditions, some portions of this selective combining might beperformed before the A to D, with others left until after the conversionto digital. In some cases, this can be done by merely reconnecting usingswitched-capacitor techniques, the capacitors of FIG. 4, duringdifferent phase sectors of the four phase sector clock. This combinesmultiple phase sector accumulated values before the A to D conversion.

One novel and advantageous case of this, an alternate value combiningsystem, is to again selectively combine the four phase sector correlatedanalog accumulated values, but now to obtain two analog accumulatedvalues to be A to D converted for each local clock cycle, the chargevalues Q1=Qb−Qd, and Q2=Qa−Qc. This is particularly advantageous iffrequency content slower than the desired signal were present as a largeinterfering signal. In this way the large interfering signal can becanceled, before it uses up range of the A to D converter 16. This ismost directly accomplished using switched-capacitor type techniques, byreconnecting the integrating capacitors of FIG. 4, or perhaps byconnecting the two non adjacent phase sectors to be combined, both tothe charge responding A to D simultaneously. This option reduces the Ato D conversion rate to two values per cycle, while also maintaining thebenefits of post conversion mixing for the I and Q reconstructedmodulation signals. This is another alternate enhancement of the currentinvention and is achieved by arranging clocking and switching of thecapacitors of FIG. 4 so that two of the charging phases are combined andonly two complete charge integrations are accomplished per full cycle ofthe clock, with each being a combination of two non-adjacent quadraturephase sectors of the full cycle clock. The I and Q signals can then bepost conversion mixed using the digital modulation signal valuesobtained from performing the A to D conversion on Q1 and Q2, and wherenow Qip=Q1+Q2=Qa+Qb−Qc−Qd, and Qq=Q1−Q2=Qb+Qc−Qa−Qd.

This alternate combining system is one of several ways the RF2D 4 can besetup to accumulate and operate on the modulated signal, the accumulatedvalues, and the modulation signal values. Different setups are likely toyield better signal to noise ratio than others, depending on signalconditions. These signal conditions can include a large set a variables,strong or weak signal, whether or not there is a strong interferingsignal nearby in frequency or physically near the receiver so that it isoverwhelming the desired signal, or whether or not there is rapidlychanging fading conditions, just to name a few. The program running inthe DSP 10 can adjust the selection of setups, to get the best signal tonoise ratio, under current signal conditions.

As previously mentioned, it is intended here that there be no timeduring which more than one of the four switches SWa, SWb, SWc, SWd aresimultaneously on. Given the imperfections of real timing andvariations, it is difficult to avoid any simultaneous on time, withoutalso providing for a period of time, however short, when all of theswitches SWa, SWb, SWc, SWd are intentionally off. This would of coursecreate a period of time, however short, where the input signal isignored. This can lead directly to aliasing. One way of overcoming thisis to make use of parasitic capacitance at the inverting input 30 of theop-amp 24. During any period where all of the switches are off, atransitory period, the parasitic capacitance at the inverting input ofthe op-amp 24, representing an auxiliary capacitive device, will beginto accumulate charge, and allow the voltage at the inverting input 30 torise slightly. The charging of this auxiliary capacitive devicecontinues until the next phase sector begins. As long as the next phasesector switch SWa, SWb, SWc, SWd turns on before this voltage rises toofar, the voltage would be pulled right back to within its properoperating range as soon as the next switch turns on, while the chargethat was accumulated by the parasitic capacitor at the inverting inputwould be redistributed exactly, onto the integrating capacitor Ca, Cb,Cc, Cd whose switch SWa, SWb, SWc, SWd is now on. In this way, after theswitching transients have settled, the now on capacitor Ca, Cb, Cc, Cdwould hold the same charge as if its switch SWa, SWb, SWc, SWd had beenturned on fully at exactly the instant the previous switch SWa, SWb,SWc, SWd had been turned fully off.

Operating a Phase Sector Integration Capturing system in this way,assuring that there is no period of time during which the input signalis ignored by subsequent processing, is an advantage referred to asContinuous Signal Capturing. One major reason why this is soadvantageous is that this completely avoids any need for anti-aliasingfilters, and avoids the complexities of managing adjustableanti-aliasing filters otherwise required for a fully flexible receiver.Continuous Signal Capturing is achieved by performing continuousaccumulation on the input signal. Whenever the input signal is no longeraccumulated on the most recently charging integration capacitor Ca, Cb,Cc, Cd, it is now effectively becomes accumulated on the subsequentphase sector's capacitor Ca, Cb, Cc, Cd. Using another circuitarchitecture might require using one or more capacitive devices devotedto this auxiliary capacitive device's purpose, but with the Phase SectorDistributed Integrator circuit configuration, this parasitic inputcapacitance at the non-inverting input 26 of the op-amp 24 accomplishesthis added feature, without the addition of a device devoted to thispurpose. There are also other ways of accomplishing continuous signalcapturing, though they are more complex in nature. This however, doesnot detract from the novelty of this entire methodology.

Another valuable enhancement to Phase Sector Integration Capturing,novel to Phase Sector Integration Capturing, is the fact that byselecting to accumulate the modulated signal input over multiple cyclesof the local clock, a Phase Sector Distributed Integrator block canaccumulate phase sector correlated values for phase sectors, A, B, C,and D, which represent low pass filtered, or decimation filteredaccumulated values. In this way, the charge stored on capacitor Ca getsadded to, by additional charge during a second phase A of a subsequentfull cycle, and by each phase A of multiple subsequent full cycles. Thecharge already on capacitor Cb gets added to by a second and multiplesubsequent phase B portions of second or multiple full cycles of the setof four switches. Capacitors Cc and Cd also, can thereby integratecharge over multiple full cycles of the set of four switches SWa, SWb,SWc, SWd. Charges accumulated in this way are still phase sectorcorrelated, with each capacitor Ca, Cb, Cc, Cd accumulating charge onlyduring its active phase sector of each local clock cycle. This issimilar to the low pass filter used in conjunction with the impulsesampling method previously described, the simplest of which just adds upthe input samples. However, with this integrating block, there is noadditional circuitry required, and no additional components beingclocked at the fastest clock rate, effectively 4 times the local clockrate. The same integrating capacitors Ca, Cb, Cc, Cd just continue toadd up the charge over multiple local clock cycles. Just as with theimpulse sampled input, the signal adds up over multiple local clockcycles, and the resulting integrated charge can be made available to asubsequent A to D converter 16 at a much slower, down-converted rate.There is also the same improvement in signal to noise ratio resultingfrom the averaging of the phase sector correlated charge components overtime, as they are integrated onto each of their respective capacitorsCa, Cb, Cc, Cd, linearly compounding the amount of charge that is inresponse to the signal, while adding only stochastically, the amount ofcharge that is present on the each capacitor Ca, Cb, Cc, Cd due tonoise.

While band narrowing does occur as a result of integrating the inputsignal over multiple full cycles of A, B, C, and D phases, this bandnarrowing is a function of the down-conversion factor, which is justFclk/f_RF2Dout, and which is under the control of the DSP 10. As before,this down-conversion factor can be chosen so as to not narrow the bandof the information contained in the modulated signal, or it can bechosen to narrow the band, if so desired.

Subsequently, both I and Q are first formed after digitization hasoccurred using a single A to D block 16. I and Q are formed in the DSP10 by combining, in the right polarity, the values collected during eachof the quadrature phase sectors. This again constitutes Post ConversionMixing with all of the same advantages as previously described, allprofound in several ways. The mixing and filtering to provide thecontrol voltage for the voltage controlled oscillator portion of the PLLis also the same as previously described.

Various methods of reconstructing a modulation signal have beendiscussed here. Each of the methods discussed has some combination of RFenvironment or receiving conditions under which it might represent thebest method of signal reception. This PSB-SDR system is capable ofswitching between all of these methods as signals are being received.This switching can be done in some cases by changing the signalsclocking various switched-capacitor circuits, in some cases by switchingdifferent components in or out of the circuits, and in many cases bymerely altering the program steps that are being applied to thedigitized signals as they propagate through the DSP block. These choicescan all be made under program control, by any programmable control unit,which most typically is just the DSP block. This can be done in responseto a modulation signal evaluator, which is most likely constituted bylines of code in the program running on the DSP, but might also beimplemented in some cases, by dedicated hardware circuitry. In eithercase, this modulation signal evaluator can evaluate the reconstructedmodulation signal relative to any number of characterization metrics,many determined by mere mathematical processing of the reconstructedmodulation signal, which could be done by any processing system, butgenerally for this PSB-SDR system, the processing system is the DSPblock.

Everything described to this point has mostly been describing thevarious features of this PSB-SDR system 2 as applied to achieve areceiver, receiving a modulated signal. The transmitter portion of aradio system can also be greatly enhanced by all of the previouslydescribed techniques. A number of conventional transmitter methods andtechniques can be applied to this system, while managed from the DSPblock 10 of this PSB-SDR 2. This system provides for a relativelywide-band, and thereby a relatively fast, acquisition of reconstructed Iand Q modulation signal values from a modulated signal. This enables theuse of a closed loop feedback system, which is amply stable,incorporating the transmitter block in the loop. For transmit mode, atransmit/receive mode control, generally a control signal or control bitfrom the DSP block 10, switches the modulated signal input to the PSCCblock 14 from a received signal over to an attenuated version of thetransmitter's output. The PSCC block 14 then provides reasonably quickfeedback to the DSP block 10, so that it can compare the actual transmitsignal to a desired transmit signal and make calculated adjustments tothe signals driving the transmit block 32, as necessary to achieve thedesired transmit signal at the output of the transmitter 32. In thisway, the output of the transmitter 32 is in effect regulated to matchthe desired signal, so that the desired transmit signal is achieved atthe output of the transmitter 32, greatly mitigating non-linearities,and thermal non-idealities, of the transmit block 32. Conventionalsystems have not been able to take this approach, because conventionalsystems have much too much delay in any signal path which reconstructs amodulation signal or component from a modulated signal by conventionalmeans.

The digital transmission values that are output from the DSP 10 areconverted to analog values by the RF DAC 34. This DAC 34 also includes afast Phase Sector Distributed Output block, which steps through variousanalog phase sector correlated output values, providing each value inthe form of either a current or a voltage proportional to this phasesector correlated output value to the transmitter 32 for transmission.The values that are distributed for each phase sector are generallyupdated much more slowly, by a digital to analog conversion of the dataprovided by the DSP 10. Because of this arrangement, the data updaterate out of the DSP 10 does not have a full RF transmission rate, butcan have an update rate more on the order of the bandwidth of theresulting RF transmission, rather than up at the carrier frequency.

This PSB-SDR system 2 provides for a zero IF, direct RF down-conversiontechnique, which converts as directly as possible, an RFband-constrained signal located at a center frequency, to a clockedparallel data stream. The center frequency at which RF information isdown-converted from is determined by the frequency of the local clock,Fclk. This local clock is developed by the local clock generation block12, under the control of the DSP 10. A down-conversion factor is givenby the ratio of the local clock to the frequency of the complete cycleoutput rate of the A to D conversion block 16, or Fclk/f_RF2Dout. Thisratio is an integer ratio, also controlled by the DSP block 10. Oneblock within the PSB-SDR 2, the RF2D 14, essentially performs an RF todigital conversion. The output of this block 4 is a stream of multiplebit wide data, provided to the DSP block 10. The PSB-SDR 2 incorporatesa transmitter 32, also controlled by the DSP 10, with a reconstructedmodulation signal generated by the RF2D block 4 from the output of thetransmitter fed back to the DSP 10, allowing the DSP 10 to maintainclosed loop control of the transmitted output signal. This system iscapable of receiving and transmitting in accordance with anytransmission or wireless standard, requiring only programming to do so,and is limited in RF application only by the operational bandwidthlimitations imposed by the semiconductor process into which it isfabricated.

FIGS. 5 a-13 are flow charts representing steps of various embodimentsof aspects of the present invention. Although the steps represented inthese Figures are presented in a specific order, the technologypresented herein can be performed in any variation of this order.Furthermore, additional steps may be executed between the stepsillustrated in these Figures. Although many of the following describedenhancements are equally applicable to PSIS and PSIC, for simplicity thedescriptions will use only the “accumulated value” terminology of PSIC.The PSIS terminology “sampled value” may be directly substituted for“accumulated value” in those descriptions to apply them to the PSISapproach.

FIGS. 5 a-b are a flow chart representing steps of one embodiment methodfor extracting values representative of modulation signal componentsfrom a modulated signal. A local clock signal is developed 50, whichcorrelates in time to the modulated signal and has a plurality ofnon-overlapping phase sectors per cycle.

The modulated signal is accumulated 52 into an accumulated value,separately for at least one phase sector of one cycle of the local clocksignal. In one embodiment, the modulated signal is accumulated by analogmeans and the accumulated values are analog accumulated values. Eachaccumulated value is representative of a modulation signal component.The modulated signal accumulated is of an amount representative of themathematical integral of the modulated signal during each phase sectorof the local clock over which the modulated signal is accumulated.

In one embodiment, each accumulated value represents the amountaccumulated over one phase sector. In an alternative embodiment, atleast one accumulated value includes amounts accumulated in eitherpolarity, selectively, over multiple phase sectors of one cycle of thelocal clock.

FIG. 6 illustrates one embodiment for accumulating the modulated signalinto an accumulated value. A capacitive device is charged 80 with themodulated signal such that the accumulated charge on the capacitivedevice is the accumulated value. The charging process 80 is repeated 82for as many clock cycles and phase sectors as desired. The accumulatedvalue is then transferred 84 to an analog to digital converter and thecapacitive device is reinitialized 86.

Returning to FIGS. 5 a-b, the accumulating step is repeated 54 duringmultiple cycles of the local clock. In one embodiment, at least oneaccumulated value includes amounts accumulated during multiple cycles ofthe local clock. In one embodiment, the accumulated values includeamounts accumulated during the same phase sector of multiple cycles ofthe local clock.

In one embodiment, accumulating the modulated signal into an accumulatedvalue includes a switched capacitor filter accumulating the modulatedsignal over multiple clock cycles. In one embodiment, the switchedcapacitor filter is a low pass filter.

In one embodiment, the accumulated values are selectively combined 58,64 to obtain at least one modulation signal value representative of themodulation signal. Selectively combining 58, 64 the accumulated valuesincludes selecting whether to combine any accumulated values, whichaccumulated values to combine, and whether to add or subtract theaccumulated values. In one embodiment, analog accumulated values areselectively combined 58 to obtain analog modulation signal valuesrepresentative of the modulation signal and the analog modulation signalvalues are digitized 60 to obtain digital modulation signal valuesrepresentative of the modulation signal.

In an alternative embodiment, analog accumulated values are digitized 62to obtain digitized accumulated values and the digitized accumulatedvalues are selectively combined 64 to obtain digital modulation signalvalues representative of the modulation signal.

The accumulating and combining steps are repeated 66 over multiplecycles of the local clock signal. In one embodiment, the resultingmodulation signal values are selectively combined 68 to reconstruct themodulation signal. Selectively combining 68 the modulation signal valuesincludes selecting whether to combine any modulation signal values,which modulation signal values to combine, and whether to add orsubtract the modulation signal values.

In one embodiment, signal conditions of the reconstructed modulationsignal are evaluated 70. The manner of selectively combining 68 theresulting modulation signal values is altered 72, as necessary ordesirable, based on the signal conditions.

In one embodiment, the accumulated values are evaluated 74 and themanner of selectively combining 58, 64 the accumulated values is altered76 based on the evaluation 74. In one embodiment, the accumulated valuesare selectively combined 58, 64 as directed by an instruction from aprogrammable control unit, of which the digital signal processor 10 isone example.

In another embodiment, the manner of developing 50 the local clocksignal is altered 78 based on the evaluation 74. Altering 78 the mannerof developing the local clock signal includes selectively altering thenumber of phase sectors per cycle, frequency, and phase of the localclock signal.

FIG. 7 illustrates one embodiment where the modulation signal includesin-phase and quadrature signals and the accumulated values form in-phaseand quadrature signal components or are combined to form in-phase andquadrature signal components. There are four phase sectors per cycle.

The modulated signal is accumulated 88, 90, 92, 94 separately duringeach phase sector of one cycle of the local clock signal, into first,second, third, and fourth accumulated values. The first, second, third,and fourth values are selectively combined 96 to obtain an in-phasevalue representative of the in-phase signal. The first, second, third,and fourth values are also selectively combined 98 to obtain aquadrature value representative of the quadrature signal.

FIG. 8 shows an alternative embodiment for obtaining the in-phase andquadrature values. The first and third values are combined 100 and thesecond and fourth values are combined 102. In one embodiment, combining100 the first and third values includes during the first and third phasesectors of the local clock signal charging a first capacitive devicewith the modulated signal and combining 102 the second and fourth valuesincludes during the second and fourth phase sectors of the local clocksignal charging a second capacitive device with the modulated signal.

The combined first and third values are digitized 104 and the combinedsecond and fourth values are digitized 106. The digitized combinedsecond and fourth values are combined 108 with the digitized combinedfirst and third values to obtain the in-phase values and the digitizedcombined second and fourth values are combined 110 with the digitizedcombined first and third values to obtain the quadrature values.

FIG. 9 illustrates another embodiment for accumulating the modulatedsignal where there are four phase sectors per cycle. During the firstphase sector of the local clock signal, a first capacitive device ischarged 112 with the modulated signal. During the second phase sector ofthe local clock signal, a second capacitive device is charged 114 withthe modulated signal. During the third phase sector of the local clocksignal, a third capacitive device is charged 116 with the modulatedsignal. During the fourth phase sector of the local clock signal, afourth capacitive device is charged 118 with the modulated signal.

In one embodiment, the modulated signal is continuously accumulated overa plurality of cycles of the local clock so that there are no intervalsduring the plurality of cycles wherein the modulated signal is notaccumulated. FIG. 10 illustrates one embodiment for continuouslyaccumulating the modulated signal. For the first accumulated value, afirst capacitive device is charged 120 with the modulated signal. Anauxiliary capacitive device is charged 122 with the modulated signalduring the transitory period between phase sectors. For the secondaccumulated value, a second capacitive device is charged 124 with themodulated signal. The accumulated charge on the auxiliary capacitivedevice is transferred 126 to the second capacitive device while thesecond capacitive device is charging 124.

The auxiliary capacitive device is charged 128 with the modulated signalduring the transitory period between charging the second capacitivedevice and the third capacitive device. The third capacitive device ischarged 130 with the modulated signal. The accumulated charge on theauxiliary capacitive device is transferred 132 to the third capacitivedevice while the third capacitive device is charging 130.

The auxiliary capacitive device is charged 134 with the modulated signalduring the transitory period between charging the third capacitivedevice and the fourth capacitive device. The fourth capacitive device ischarged 136 with the modulated signal. The accumulated charge on theauxiliary capacitive device is transferred 138 to the fourth capacitivedevice while the fourth capacitive device is charging 138.

The auxiliary capacitive device is charged 140 with the modulated signalduring the transitory period between charging the fourth capacitivedevice and the first capacitive device. The first capacitive device ischarged 120 with the modulated signal. The accumulated charge on theauxiliary capacitive device is transferred 142 to the first capacitivedevice while the first capacitive device is charging 120. Although thepreceding description refers to only one auxiliary capacitive device,multiple auxiliary capacitive devices may be used to accomplish thiscapture and transfer of charge accumulated between phase sectors.

FIG. 11 illustrates an alternate embodiment for accumulating themodulated signal into an accumulated value. A first accumulation blockis activated 144. When activated, the first accumulation blockaccumulates 146 the modulated signal into an accumulated value. Whilethe first accumulation block is accumulating 146, an accumulated valueis transferred 148 from a second accumulation block. The secondaccumulation block is activated 150. When activated, the secondaccumulation block accumulates 152 the modulated signal into anaccumulated value. While the second accumulation block is accumulating152, an accumulated value is transferred 154 from the first accumulationblock.

FIG. 12 illustrates one embodiment for the method of the presentinvention including a transmit mode. The present invention is useful inboth a receive mode and a transmit mode and may be selectively switched156 between the modes. In one embodiment, a single digital signalprocessing system selectively switches 156 between the transmit andreceive modes, supplying data to a transmitter to transmit, andobtaining data from a receiver.

In the receive mode, the modulated signal is a received 158 modulatedsignal. In the transmit mode, a transmit signal is generated 160 and themodulated signal is generated 162 by attenuating the transmit signal. Inthe transmit mode, a desired transmission modulation signal is compared164 to a selectable combination of the accumulated values. Thegeneration 160 of the transmit signal is regulated 164 with thecomparison between the desired transmission modulation signal and theselectable combination of the accumulated values.

FIG. 13 illustrates an alternate embodiment for altering the local clocksignal. The accumulated values are selectively combined 166 to obtain atleast one control signal value representative of a control signal. Theaccumulating and combining steps are repeated 168 over multiple cyclesof the local clock signal. The resulting control signal values areselectively combined 170 to construct a control signal. Signalconditions of the control signal are evaluated 172. The manner ofdeveloping the local clock signal is altered 174 based on the signalconditions.

The methods and systems disclosed herein accomplish the highest goal ofSDR, to provide hardware which provides fully flexible reception andtransmission, in any programmable format, of any modulated signal, andwithout compromising performance.

The foregoing description is only illustrative of the invention. Variousalternatives and modifications can be devised by those skilled in theart without departing from the invention. Accordingly, the presentinvention embraces all such alternatives, modifications, and variancesthat fall within the scope of the appended claims.

What is claimed is:
 1. A method for extracting values representative ofmodulation signal components from a modulated signal, the modulatedsignal containing a modulation signal, the method comprising: developinga local clock signal which correlates in time to the modulated signaland has a plurality of non-overlapping phase sectors per cycle;acquiring at least one signal value from the modulated signal,separately for at least one phase sector of a plurality of cycles of thelocal clock signal, wherein acquiring is selected from the groupconsisting of sampling, impulse sampling, capturing, accumulating, andintegrating; and combining at least two signal values from the samephase sector, but different local clock cycles to obtain at least onecombined signal value representative of a modulation signal component.2. The method of claim 1 wherein: acquiring at least one signal valueincludes sampling the modulated signal to acquire at least one sampledvalue and combining at least two signal values from the same phasesector, but different clock cycles to obtain at least one combinedsignal value representative of the modulation signal component includescombining at least two sampled values from the same phase sector, butdifferent clock cycles to obtain at least one combined sampled valuerepresentative of a modulation signal component.
 3. The method of claim2 wherein sampling the modulated signal includes, for each sampledvalue, charging a sampling capacitive device with the modulated signalsuch that the charge accumulated on the sampling capacitive device isthe signal value.
 4. The method of claim 3 wherein combining the atleast two sampled values from the same phase sector includes charging acombining capacitive device with the charge from each samplingcapacitive device such that the accumulated charge on the combiningcapacitive device is the combined sampled value.
 5. The method of claim1 wherein combining at least two signal values from the same phasesector, but different clock cycles, includes low pass filteringseparately for each phase sector, signal values acquired during thatphase sector over multiple clock cycles.
 6. The method of claim 1wherein the number of local clock cycles over which the signal valuesare acquired is variable to achieve a phase sector correlated decimationfilter having a variable decimation rate.
 7. The method of claim 1wherein combining at least two signal values from the same phase sector,but different clock cycles, includes accumulating at least two signalvalues from the same phase sector, but different clock cycles.
 8. Themethod of claim 1 further including: evaluating the combined signalvalues and altering a manner of developing the local clock signal basedon the evaluation.
 9. The method of claim 1 further including:selectively combining the combined signal values to obtain at least onecontrol signal value representative of a control signal; repeating theacquiring and combining steps over multiple cycles of the local clocksignal; selectively combining the resulting control signal values toconstruct the control signal; evaluating signal conditions of thecontrol signal; and altering a manner of developing the local clocksignal based on the signal conditions.
 10. The method of claim 1 whereinthe signal values are analog signal values and the combined signal valueis an analog combined signal value and further including digitizing theanalog combined signal value to obtain a digitized combined signalvalue.
 11. The method of claim 1 wherein the combined signal values formin-phase and quadrature signal components.
 12. The method of claim 1further including combining the combined signal values to form in-phaseand quadrature signal components.
 13. The method of claim 1 furtherincluding receiving the modulated signal and wherein the modulatedsignal is a received modulated signal.
 14. The method of claim 1 furtherincluding: in a receive mode, receiving the modulated signal and whereinthe modulated signal is a received modulated signal, and in a transmitmode, generating a transmit signal and the modulated signal, as anattenuated version of the transmit signal, generating the transmitsignal and the modulated signal regulated by a comparison between adesired transmission modulation signal and a selectable combination ofthe combined signal values; and selectively switching between thetransmit and receive modes.
 15. The method of claim 1 wherein acquiringthe modulated signal includes a switched capacitor filter acquiring themodulated signal over multiple clock cycles.
 16. A system for extractingvalues representative of modulation signal components from a modulatedsignal, the modulated signal containing a modulation signal, the systemcomprising: a timing system configured to develop a local clock signalwhich correlates in time to the modulated signal, and has a plurality ofnon-overlapping phase sectors per cycle; a phase sector correlatedcapturing block configured to acquire at least one signal value from themodulated signal, separately for at least one phase sector of aplurality of cycles of the local clock signal, wherein acquiring isselected from the group consisting of sampling, impulse sampling,capturing, accumulating, and integrating; and a signal value combiningsystem configured to combine at least two signal values from the samephase sector, but different clock cycles to obtain at least one combinedsignal value representative of a modulation signal component.
 17. Thesystem of claim 16 wherein: acquiring at least one signal value includessampling the modulated signal to acquire at least one sampled value andcombining at least two signal values from the same phase sector, butdifferent clock cycles to obtain at least one combined signal valuerepresentative of a modulation signal component includes combining atleast two sampled values from the same phase sector, but different clockcycles to obtain at least one combined sampled value representative of amodulation signal component.
 18. The system of claim 17 furtherincluding at least one capacitive device and wherein the phase sectorcorrelated capturing block is further configured to, for each sampledvalue, charge one of the capacitive devices with the modulated signalsuch that the sampled charge on the capacitive device is the sampledvalue.
 19. The system of claim 18 further including a combiningcapacitive device and wherein the sampled value combining systemconfigured to charge the combining capacitive device with the chargefrom each sampling capacitive device such that the charge accumulated onthe combining capacitive device is the combined sampled value.
 20. Thesystem of claim 16 wherein the signal value combining system includes alow pass filter configured to low pass filter separately for each phasesector, signal values acquired during that phase sector over multipleclock cycles.
 21. The system of claim 16 wherein the number of localclock cycles over which the signal values are acquired is variable andwherein the signal value combining system includes a phase sectorcorrelated decimation filter having a variable decimation rate.
 22. Thesystem of claim 16 wherein the signal value combining system is furtherconfigure to accumulate at least two signal values from the same phasesector, but different clock cycles.
 23. The system of claim 16 whereinthe phase sector correlated capturing block is further configured torepeat the acquiring step during multiple cycles of the local clock andfurther including a modulation signal evaluator configured to evaluatethe resulting signal values and wherein the timing system is furtherconfigured to alter a manner of developing the local clock signal basedon the evaluation.
 24. The system of claim 16 wherein the phase sectorcorrelated capturing block is further configured to repeat the acquiringstep over multiple cycles of the local clock and further configured toconfigured to selectively combine the signal values to obtain at leastone control signal value representative of a control signal and torepeat the combining step over multiple cycles of the local clock signaland further including: a modulation value combining system configured toselectively combine the resulting control signal values to construct thecontrol signal; a control signal evaluator configured to evaluate signalconditions of the control signal; and wherein the timing system isfurther configured to alter a manner of developing the local clocksignal based on the signal conditions.
 25. The system of claim 16wherein the signal values are analog signal values and the combinedsignal value is an analog combined signal value and further including ananalog to digital converter configured to digitize the analog combinedsignal value to obtain a digitized combined signal value.
 26. The systemof claim 16 wherein the modulation signal values form in-phase andquadrature signal components.
 27. The system of claim 16 wherein thesignal value combining system is further configured to selectivelycombine the modulation signal values to form in-phase and quadraturesignal components.
 28. The system of claim 16 further including anantenna configured to receive the modulated signal and wherein themodulated signal is a received modulated signal.
 29. The system of claim16 further including: an antenna configured to receive the modulatedsignal and wherein the modulated signal is a received modulated signalin a receive mode, and a transmitter configured to generate a transmitsignal and the modulated signal, as an attenuated version of thetransmit signal, generating the transmit signal and the modulated signalregulated by a comparison between a desired transmission modulationsignal and a selectable combination of the signal values, in a transmitmode, and a transmit/receive mode control configured to switch betweenthe transmit and receive modes.
 30. The system of claim 16 wherein thephase sector correlated capturing block is further configured to repeatthe acquiring step during multiple cycles of the local clock and furtherincludes a switched capacitor filter configured to accumulate themodulated signal over multiple clock cycles.